In order to provide the widest possible coverage for a digital transmission, such as for cell phones or a digital television broadcast, it's desirable to use multiple transmitters that are separated from each other spatially. This permits a wider area to be covered, uses less total broadcast power, and can help to fill in dark areas where the transmission from one transmitter may be blocked. Thus, using multiple transmitters can provide wider and more complete coverage for virtually any digital transmission.
However, using multiple transmitters creates a serious problem when the receiver is at a “seam” between two transmitters, because the additional signal can appear as a “ghost” that can be as large as the “main” signal. Furthermore, destructive interference creates a series of perfect or near perfect nulls.
Existing receiver technology handles ghosts by filtering them out in order to interpret the “main” signal. But in a multi-transmitter environment this strategy is unworkable. It makes little sense to design a system to filter out a ghost that can be an arbitrarily large fraction of the “main” signal's size. Furthermore, near the margins the best this subtractive strategy can ever provide is a signal strength equal to the stronger transmitter's signal—the energy from the secondary signal is wasted.
Even when the ghosts are smaller than 100% of the “main” signal, there is an equal probability of pre- and post-ghosts. In the most common situation, the strongest signal is the one following the most direct path. Ghosts are most often produced by “multipathing,” that is, by portions of the signal following paths of different lengths from the transmitter to the receiver. Thus, ghosts are typically produced by one or more strong reflections. The first signal to arrive is typically the most direct, and therefore the strongest, and so in the usual situation the ghost is a post-ghost. In a multi-transmitter environment, though, while the receiver is near a seam the stronger signal can easily arrive after the ghost. With signals arriving from two directions, it is possible that the more direct path may be the longer one. Consequently, pre-ghosts are about as likely as post-ghosts, and may be arbitrarily strong. Furthermore, if the transmitters are out of sync with each other by even a small amount, where the one lagging happens to be the closer one the receiver will likely see pre-ghosts.
Existing technology relies on the assumption that post-ghosts predominate (i.e., existing systems are not generally designed to deal with Raleigh fading). Thus, existing receivers generally will be either inefficient or incapable of dealing with a multi-transmitter environment, even if the ghosts are sufficiently small compared to the “main” signal.
In short, in a multi-transmitter environment, the “main” signal becomes a meaningless concept at the seams of the transmission. In order to operate efficiently in a multi-transmitter environment, a digital receiver must operate with a different paradigm. What is needed is a digital receiver that employs an additive strategy—that is, one in which the energy from one or more relatively large ghosts can be captured and used to aid in the synchronization process, rather than filtered out and discarded. Such a receiver could both function with ghosts 100% of the size of the “main” signal, and provides substantially superior performance whenever ghosts exceed about 70% of the size of the “main” signal.
From the receiver's perspective, most of the signal is useless for synchronization, because it is indistinguishable from white noise. The more information that is packed into a signal, the more closely it will resemble white noise, so this is both a desirable and inevitable feature of the signal. Nevertheless, some bandwidth must be “wasted” in order to provide the receiver a means to orient itself. Typically, one of two strategies is employed. In some systems, a pilot signal is included. This is a sharp peak of energy in a very narrow frequency band, which is very easy for the receiver to pick out.
A phase-lock loop, such as the one shown in FIG. 1, indicated generally at 100, is a typical way to synch up a receiver using a pilot. A multiplier 110 multiplies the signal and the output of a voltage controlled oscillator 120 (“VCO”) to produce a beat note (a sine wave with a frequency equal to the difference between the frequency of the pilot signal and the VCO's output). The beat note passes through a low-pass filter 130. The output of the filter 130 is amplified and input to the VCO 120 to complete the feedback loop. The low-pass filter 130 has competing design parameters. The more narrow the band pass of the filter 130 the smaller the response, so the slower the loop 100 is to lock up. However, a wide pass filter passes more noise and makes it harder for the loop 100 to capture at all.
It will be appreciated that the response of the loop 100 is driven by the frequency difference output of the first multiplier 110. The direction of error can only be determined by observing the slope of the time rate of change of the output. The second filter 130 distorts the sine wave, increasing the amplitude on the closer side, and decreasing it on the further side. Convergence is driven by this asymmetry of the distorted beat note.
However, because the amplitude of the beat note drops with increasing frequency difference, that distortion output drops as well, so the response of the phase-lock loop 100 decreases as the frequency of the VCO 120 diverges from the signal frequency. Thus, unless the signal happens to be close to the initial VCO 120 frequency, it will converge slowly, or not at all. A typical phase lock loop can capture when the initial VCO 120 frequency is within a factor of about 3-10 times the bandwidth of the loop.
Another, more robust, strategy for synching is to provide a signal in which information in the data is redundant in the frequency domain. The receiver can look for a correlation in the data created by this repetition to synch up. The receiver could use this same technique to find correlations in the data from signals from multiple transmitters. In mathematical terms, the correlation between the repeated signal portion can be identified by fully complex convolution. Convolution inherently corrects for the asymmetry produced by the slope of the Nyquist band, so that the peak value occurs when the limits of integration exactly correspond to the beginning and the end of the repeated data segment (and it's negative time image).
A typical existing means for performing such a convolution is the Costas Loop, shown in FIG. 2. The Costas Loop operates on a complex signal, such as a QAM signal. As with the phase-lock loop, a first multiplier 210 multiplies the signal with the output of a VCO 220, though, as shown in FIG. 2, this is a complex multiplication, which produces both an I′ and a Q′ output. As with the phase-lock loop, the output of the first multiplier is passed through a low-pass filter 230 where the unwanted (frequency sum) portion of multiplied signal is removed. The in-phase and quadrature portions are then multiplied by a second multiplier 240 to produce a beat note (assuming the sideband isn't balanced—otherwise it's merely a DC voltage.) The beat note is passed through a second low-pass filter 250, then amplified at 299 and returned to the VCO 220 to complete the feedback loop. Thus, the portion of the Costas loop following the second multiplier 240, which drives the convergence of the loop, is basically a phase-lock loop. Consequently, like the phase-lock loop, the Costas loop has the disadvantage of slow convergence.
A frequency-and-phase-lock loop (“FPLL”) (shown in FIG. 3, and described in U.S. Pat. No. 4,072,909 to Citta, which is hereby incorporated by reference in its entirety) provides faster convergence. The FPLL has a first low-pass filter 330 and a second low-pass filter 350 which perform the function of the second low-pass filter 250 in the Costas loop, which separate the averaging and noise-elimination functions. Thus, the first low-pass filter 330 can have a relatively wide band pass, so that the FPLL can acquire even when the signal and initial VCO frequencies are off by as much as a factor of 1000. The second low-pass filter 350 can have a relatively narrow band-pass, in order to give good averaging during lock-up. The output of the second multiplier 340 is a rectified sine wave with a DC offset. The DC offset provides the direction information, rather than an integration of a distorted sine wave, which provides a much stronger response when the frequency difference is relatively large. The signal from the filter 350 is amplified at 399 and returned to the VCO 320 to complete the feedback loop.
Because of the way the FPLL uses the complex information to provide both magnitude and direction information, it locks up faster, and phase noise that is less than 90 degrees out of phase doesn't disrupt the lock. However, the FPLL does not perform a convolution of the data, and is therefore dependent upon a pilot to operate. It is therefore not suitable for use with, for example, a double sideband suppressed signal.
Because of the way the FPLL uses the complex information to provide both magnitude and direction information, it locks up faster, and phase noise that is less than 90 degrees out of phase doesn't disrupt the lock. However, the FPLL does not perform a convolution of the data, and is therefore dependent upon a pilot to operate. It is therefore not suitable for use with, for example, a double sideband suppressed signal.
Thus, what is needed is a new data-synch loop which combines the desired features of the Costas Loop—synching by finding a correlation in repeated data through convolution—with the desired faster convergence of a frequency-and-phase-lock loop.